## Elementary Filter Sections

This section gives condensed analysis summaries of the four most elementary digital filters: the one-zero, one-pole, two-pole, and two-zero filters. Despite their relative simplicity, they are quite valuable to master in practice. In particular, recall from Chapter 9 that every causal, finite-order, LTI filter (any difference equation of the form Eq.(5.1)) may be factored into a series and/or parallel combinationof such sections. Implementing high-order filters as parallel and/or series combinations of low-order sections offers several advantages, such as numerical robustness and easier/safer control in real time.

### One-Zero

Figure B.1 gives the signal flow graph for the general one-zero filter. The frequency response for the one-zero filter may be found by the following steps:
By factoring out from the frequency response, to balance the exponents of , we can get this closer to polar form as follows:

We now apply the general equations given in Chapter 7 for filter gain and filter phase as a function of frequency:

A plot of and for and various real values of , is given in Fig.B.2. The filter has a zero at in the plane, which is always on the real axis. When a point on the unit circle comes close to the zero of the transfer function the filter gain at that frequency is low. Notice that one real zero can basically make either a highpass ( ) or a lowpass filter ( ). For the phase response calculation using the graphical method, it is necessary to include the pole at .

### One-Pole

Fig.B.3 gives the signal flow graph for the general one-pole filter. The road to the frequency response goes as follows:
The one-pole filter has a transfer function (hence frequency response) which is the reciprocal of that of a one-zero. The analysis is thus quite analogous. The frequency response in polar form is given by

A plot of the frequency response in polar form for and various values of is given in Fig.B.4. The filter has a pole at , in the plane (and a zero at = 0). Notice that the one-pole exhibits either a lowpass or a highpass frequency response, like the one-zero. The lowpass character occurs when the pole is near the point (dc), which happens when approaches . Conversely, the highpass nature occurs when is positive. The one-pole filter section can achieve much more drastic differences between the gain at high frequencies and the gain at low frequencies than can the one-zero filter. This difference is achieved in the one-pole by gain boost in the passband rather than attenuation in the stopband; thus it is usually desirable when using a one-pole filter to set to a small value, such as , so that the peak gain is 1 or so. When the peak gain is 1, the filter is unlikely to overflow.B.1 Finally, note that the one-pole filter is stable if and only if .

### Two-Pole

The signal flow graph for the general two-pole filter is given in Fig.B.5. We proceed as usual with the general analysis steps to obtain the following:
The numerator of is a constant, so there are no zeros other than two at the origin of the plane. The coefficients and are called the denominator coefficients, and they determine the two poles of . Using the quadratic formula, the poles are found to be located at

When the coefficients and are real (as we typically assume), the poles must be either real (when ) or form a complex conjugate pair (when ). When both poles are real, the two-pole can be analyzed simply as a cascade of two one-pole sections, as in the previous section. That is, one can multiply pointwise two magnitude plots such as Fig.B.4a, and add pointwise two phase plots such as Fig.B.4b. When the poles are complex, they can be written as

since they must form a complex-conjugate pair when and are real. We may express them in polar form as

where

is the pole radius, or distance from the origin in the -plane. As discussed in Chapter 8, we must have for stability of the two-pole filter. The angles are the poles' respective angles in the plane. The pole angle corresponds to the pole frequency via the relation

where denotes the sampling interval. See Chapter 8 for a discussion and examples of pole-zero plots in the complex -plane. If is sufficiently large (but less than 1 for stability), the filter exhibits a resonanceB.2 at radian frequency . We may call or the center frequency of the resonator. Note, however, that the resonance frequency is not usually the precise frequency of peak-gain in a two-pole resonator (see Fig.B.9 on page ). The peak of the amplitude response is usually a little different because each pole sits on the other's skirt,'' which is slanted. (See §B.1.5 and §B.6 for an elaboration of this point.) Using polar form for the (complex) poles, the two-pole transfer function can be expressed as
 (B.1)

Comparing this to the transfer function derived from the difference equation, we may identify

The difference equation can thus be rewritten as

 (B.2)

Note that coefficient depends only on the pole radius R (which determines damping) and is independent of the resonance frequency, while is a function of both. As a result, we may retune the resonance frequency of the two-pole filter section by modifying only. The gain at the resonant frequency , is found by substituting into Eq.(B.1) to get
 (B.3)

See §B.6 for details on how the resonance gain (and peak gain) can be normalized as the tuning of is varied in real time. Since the radius of both poles is , we must have for filter stability8.4). The closer is to 1, the higher the gain at the resonant frequency . If , the filter degenerates to the form , which is a nothing but a scale factor. We can say that when the two poles move to the origin of the plane, they are canceled by the two zeros there.

#### ResonatorBandwidth in Terms of Pole Radius

The magnitude of a complex pole determines the damping or bandwidth of the resonator. (Damping may be defined as the reciprocal of the bandwidth.) As derived in §8.5, when is close to 1, a reasonable definition of 3dB-bandwidth is provided by
 (B.4) (B.5)

where is the pole radius, is the bandwidth in Hertz (cycles per second), and is the sampling interval in seconds. Figure B.6 shows a family of frequency responses for the two-pole resonator obtained by setting and varying . The value of in all cases is , corresponding to . The analytic expressions for amplitude and phase response are

where and .

### Two-Zero

The signal flow graph for the general two-zero filter is given in Fig.B.7, and the derivation of frequency response is as follows:
As discussed in §5.1, the parameters and are called the numerator coefficients, and they determine the two zeros. Using the quadratic formula for finding the roots of a second-order polynomial, we find that the zeros are located at

If the zeros are real [ ], then the two-zero case reduces to two instances of our earlier analysis for the one-zero. Assuming the zeros to be complex, we may express the zeros in polar form as and , where . Forming a general two-zero transfer function in factored form gives

from which we identify and , so that

is again the difference equation of the general two-zero filter with complex zeros. The frequency , is now viewed as a notch frequency, or antiresonance frequency. The closer R is to 1, the narrower the notch centered at . The approximate relation between bandwidth and given in Eq.(B.5) for the two-pole resonator now applies to the notch width in the two-zero filter. Figure B.8 gives some two-zero frequency responses obtained by setting to 1 and varying . The value of , is again . Note that the response is exactly analogous to the two-pole resonator with notches replacing the resonant peaks. Since the plots are on a linear magnitude scale, the two-zero amplitude response appears as the reciprocal of a two-pole response. On a dB scale, the two-zero response is an upside-down two-pole response.

### Complex Resonator

Normally when we need a resonator, we think immediately of the two-pole resonator. However, there is also a complex one-pole resonator having the transfer function

 (B.6)

where is the single complex pole, and is a scale factor. In the time domain, the complex one-pole resonator is implemented as

Since is complex, the output is generally complex even when the input is real. Since the impulse response is the inverse z transform of the transfer function, we can write down the impulse response of the complex one-pole resonator by recognizing Eq.(B.6) as the closed-form sum of an infinite geometric series, yielding

where, as always, denotes the unit step function:

Thus, the impulse response is simply a scale factor times the geometric sequence with the pole as its term ratio''. In general, is a sampled, exponentially decaying sinusoid at radian frequency . By setting somewhere on the unit circle to get

we obtain a complex sinusoidal oscillator at radian frequency rad/sec. If we like, we can extract the real and imaginary parts separately to create both a sine-wave and a cosine-wave output:

These may be called phase-quadrature sinusoids, since their phases differ by 90 degrees. The phase quadrature relationship for two sinusoids means that they can be regarded as the real and imaginary parts of a complex sinusoid. By allowing to be complex,

we can arbitrarily set both the amplitude and phase of this phase-quadrature oscillator:

The frequency response of the complex one-pole resonator differs from that of the two-pole real resonator in that the resonance occurs only for one positive or negative frequency , but not both. As a result, the resonance frequency is also the frequency where the peak-gain occurs; this is only true in general for the complex one-pole resonator. In particular, the peak gain of a real two-pole filter does not occur exactly at resonance, except when , , or . See §B.6 for more on peak-gain versus resonance-gain (and how to normalize them in practice).

#### Two-PolePartial Fraction Expansion

Note that every real two-pole resonator can be broken up into a sum of two complex one-pole resonators:

 (B.7)

where and are constants (generally complex). In this parallel one-pole'' form, it can be seen that the peak gain is no longer equal to the resonance gain, since each one-pole frequency response is tilted'' near resonance by being summed with the skirt'' of the other one-pole resonator, as illustrated in Fig.B.9. This interaction between the positive- and negative-frequency poles is minimized by making the resonance sharper ( ), and by separating the pole frequencies . The greatest separation occurs when the resonance frequency is at one-fourth the sampling rate ( ). However, low-frequency resonances, which are by far the most common in audio work, suffer from significant overlapping of the positive- and negative-frequency poles.
To show Eq.(B.7) is always true, let's solve in general for and given and . Recombining the right-hand side over a common denominator and equating numerators gives

which implies

The solution is easily found to be

where we have assumed im, as necessary to have a resonator in the first place. Breaking up the two-pole real resonator into a parallel sum of two complex one-pole resonators is a simple example of a partial fraction expansion (PFE) (discussed more fully in §6.8). Note that the inverse z transform of a sum of one-pole transfer functions can be easily written down by inspection. In particular, the impulse response of the PFE of the two-pole resonator (see Eq.(B.7)) is clearly

Since is real, we must have , as we found above without assuming it. If , then is a real sinusoid created by the sum of two complex sinusoids spinning in opposite directions on the unit circle.

The term biquad'' is short for bi-quadratic'', and is a common name for a two-pole, two-zero digital filter. The transfer function of the biquad can be defined as

 (B.8)

where can be called the overall gain of the biquad. Since both the numerator and denominator of this transfer function are quadratic polynomials in (or ), the transfer function is said to be bi-quadratic'' in (or ). As derived in §B.1.3, for real second-order polynomials having complex roots, it is often convenient to express the polynomial coefficients in terms of the radius and angle of the positive-frequency pole. For example, denoting the denominator polynomial by , we have

This representation is most often used for the denominator of the biquad, and we think of as the resonance frequency (in radians per sample-- , where is the resonance frequency in Hz), and determines the Q'' of the resonance (see §B.1.3). The numerator is less often represented in this way, but when it is, we may think of the zero-angle as the antiresonance frequency, and the zero-radius affects the depth and width of the antiresonance (or notch). As discussed on page , a common setting for the zeros when making a resonator is to place one at (dc) and the other at (half the sampling rate), i.e., and in Eq.(B.8) above . This zero placement normalizes the peak gain of the resonator if it is swept using the parameter. Using the shift theorem for z transforms, the difference equation for the biquad can be written by inspection of the transfer function as

where denotes the input signal sample at time , and is the output signal. This is the form that is typically implemented in software. It is essentially the direct-form I implementation. (To obtain the official direct-form I structure, the overall gain must be not be pulled out separately, resulting in feedforward coefficients instead. See Chapter 9 for more about filter implementation forms.)

        outputsignal = filter(B,A,inputsignal);

  typedef double *pp; // pointer to array of length NTICK typedef word double; // signal and coefficient data type typedef struct _biquadVars { pp output; pp input; word s2; word s1; word gain; word a2; word a1; word b2; word b1; } biquadVars; void biquad(biquadVars *a) { int i; dbl A; word s0; for (i=0; igain * a->input[i]; A -= a->a1 * a->s1; A -= a->a2 * a->s2; s0 = A; A += a->b1 * a->s1; a->output[i] = a->b2 * a->s2 + A; a->s2 = a->s1; a->s1 = s0; } }