## FDN Reverberation

Feedback Delay Networks (FDN) were introduced earlier in §2.7. An example is shown in Fig.2.29 on page . After a brief historical summary, this section will cover some practical considerations for the use of FDNs as reverberators.

### History of FDNs for Artificial Reverberation

Feedback delay networks were first suggested for artificial reverberation by Gerzon [156], who proposed an ``orthogonal matrix feedback reverberation unit''. He noted that individual feedback comb filters yielded poor quality, but that several such filters could sound good when cross-coupled. An ``orthogonal matrix feedback'' around a parallel bank of delay lines was suggested as a means of obtaining maximally rich cross-coupling. He was especially concerned with good stereo spreading of the reverberation at a time when most artificial reverberators sought merely to decorrelate the reverberation in each output channel.

Later, and apparently independently, Stautner and Puckette [473] suggested a specific four-channel FDN reverberator and gave general stability conditions for the FDN. They proposed the feedback matrix

More recently, Jot [217,216] developed a systematic FDN design methodology allowing largely independent setting of reverberation time in different frequency bands. Using Jot's methodology, FDN reverberators can be polished to a high degree of quality, and they are presently considered to be among the best choices for high-quality artificial reverberation.

Jot's early work was concerned only with single-input, single-output
(SISO) reverberators. Later work [218] with Jullien
and others at IRCAM was concerned also with *spatializing* the
reverberation.

An example FDN reverberator using three delay lines is shown in
Fig.3.10. It can be seen as an FDN (introduced in §2.7),
plus an additional low-order filter applied to the non-direct
signal. This filter is called a ``tonal correction'' filter by Jot,
and it serves to equalize modal energy irrespective of the
reverberation time in each band. In other words, if the decay time is
made very short in some band, will have a large gain in that
band so that the total energy in the band's impulse-response is
unchanged. This is another example of *orthogonalization* of
reverberation parameters: In this case, adjustments in reverberation
time, in any frequency band, do not alter total signal energy in the
impulse response in that band.

### Choice of Lossless Feedback Matrix

As mentioned in §3.4, an ``ideal'' late reverberation
impulse response should resemble exponentially decaying noise
[314]. It is therefore useful when designing a
reverberator to start with an infinite reverberation time (the
``lossless case'') and work on making the reverberator a good ``noise
generator''. Such a starting point is ofen referred to as a
*lossless prototype*
[153,430]. Once
smooth noise is heard in the impulse response of the lossless
prototype, one can then work on obtaining the desired reverberation
time in each frequency band (as will be discussed in §3.7.4
below).

In reverberators based on feedback delay networks (FDNs), the smoothness of the ``perceptually white noise'' generated by the impulse response of the lossless prototype is strongly affected by the choice of FDN feedback matrix as well as the (ideally mutually prime) delay-line lengths in the FDN (discussed further in §3.7.3 below). Following are some of the better known feedback-matrix choices.

#### Hadamard Matrix

A second-order *Hadamard matrix* may be defined by

*e.g.*,

*mixing and scattering*property of the matrix.

As of version 0.9.30, Faust's `math.lib`^{4.12}contains a function called `hadamard(n)` for generating an
Hadamard matrix, where must be a power of . A
Hadamard feedback matrix is used in the programming example
`reverb_designer.dsp` (a configurable FDN reverberator)
distributed with Faust.

A Hadamard feedback matrix is said to be used in the IRCAM Spatialisateur [218].

#### Householder Feedback Matrix

One choice of lossless feedback matrix
for FDNs, especially
nice in the case, is a specific *Householder
reflection* proposed by Jot [217]:

where can be interpreted as the specific vector about which the input vector is reflected in -dimensional space (followed by a sign inversion). More generally, the identity matrix can be replaced by any permutation matrix [153, p. 126].

It is interesting to note that when is a power of 2, no multiplies are required [430]. For other , only one multiply is required (by ).

Another interesting property of the Householder reflection
given by Eq.(3.4) (and its permuted forms) is that an
matrix-times-vector operation may be carried out with only
additions (by first forming times the input vector, applying
the scale factor , and subtracting the result from the input
vector). This is the same computation as physical *wave
scattering* at a junction of identical waveguides (§C.8).

An example implementation of a Householder FDN for is shown in Fig.3.11. As observed by Jot [153, p. 216], this computation is equivalent to parallel feedback comb filters with one new feedback path from the output to the input through a gain of .

A nice feature of the Householder feedback matrix is that
for , *all entries in the matrix are nonzero*. This
means every delay line feeds back to every other delay line, thereby
helping to maximize echo density as soon as possible.

Furthermore, for , all matrix entries have the *same
magnitude*:

*decoupled*parallel comb filters.

Due to the elegant balance of the Householder feedback matrix, Jot [216] proposes an FDN based on an embedding of feedback matrices:

#### Householder Reflections

For completeness, this section derives the Householder reflection
matrix from geometric considerations [451]. Let
denote
the *projection matrix* which orthogonally projects vectors onto
, *i.e.*,

*difference vector*between and , its orthogonal projection onto , since

*minus*this difference vector gives a

*reflection*of the vector about :

*reflecting*about --a so-called

*Householder reflection*.

#### Most General Lossless Feedback Matrices

As shown in §C.15.3, an FDN feedback matrix is lossless if and only if its eigenvalues have modulus 1 and its eigenvectors are linearly independent.

A *unitary matrix* is any (complex) matrix that is inverted
by its own (conjugate) transpose:

*Hermitian conjugate*(

*i.e.*, the complex-conjugate transpose) of . When is real (as opposed to complex), we may simply call it an

*orthogonal matrix*, and we write , where denotes matrix transposition.

All unitary (and orthogonal) matrices have unit-modulus eigenvalues and linearly independent eigenvectors. As a result, when used as a feedback matrix in an FDN, the resulting FDN will be lossless (until the delay-line damping filters are inserted, as discussed in §3.7.4 below).

#### Triangular Feedback Matrices

An interesting class of feedback matrices, also explored by Jot
[216], is that of *triangular
matrices*. A basic fact from linear algebra
is that triangular matrices (either lower or upper triangular) have
all of their eigenvalues along the diagonal.^{4.13} For example, the
matrix

It is important to note that not all triangular matrices are lossless. For example, consider

One way to avoid ``coupled repeated poles'' of this nature is to use non-repeating eigenvalues. Another is to convert to Jordan canonical form by means of a similarity transformation, zero any off-diagonal elements, and transform back [329].

### Choice of Delay Lengths

Following Schroeder's original insight, the delay line lengths in an
FDN ( in Fig.3.10) are typically chosen to be *mutually
prime*. That is, their prime factorizations contain no common
factors. This rule maximizes the number of samples that the lossless
reverberator prototype must be run before the impulse response
repeats.

The delay lengths should be chosen to ensure a
*sufficiently high mode density* in all frequency bands. An
insufficient mode density can be heard as ``ringing tones'' or an
uneven amplitude modulation in the late reverberation impulse
response.

#### Mean Free Path

A rough guide to the average delay-line length is the ``mean free
path'' in the desired reverberant environment. The *mean free
path* is defined as the average distance a ray of sound travels before
it encounters an obstacle and reflects. An approximate value for the
mean free path, due to Sabine, an early pioneer of statistical room
acoustics, is

*diffuse field*assumption,

*i.e.*, that plane waves are traveling randomly in all directions [349,47] (see §3.2.1 for a simple construction). Normally, late reverberation satisfies this assumption well, away from open doors and windows, provided the room is not too ``dead''. Regarding each delay line as a mean-free-path delay, the average can be set to the mean free path by equating

*diffuse*, especially at high frequencies. In a diffuse reflection, a single incident plane wave reflects in many directions at once.

#### Mode Density Requirement

A guide for the *sum* of the delay-line lengths is the desired
*mode density*. The sum of delay-line lengths in a lossless
FDN is simply the *order* of the system :

Since the order of a system equals the number of poles, we have that is the number of poles on the unit circle in the lossless prototype. If the modes were uniformly distributed, the mode density would be modes per Hz. Schroeder [417] suggests that, for a reverberation time of 1 second, a mode density of 0.15 modes per Hz is adequate. Since the mode widths are inversely proportional to reverberation time, the mode density for a reverberation time of 2 seconds should be 0.3 modes per Hz, etc. In summary, for a sufficient mode density in the frequency domain, Schroeder's formula is

#### Prime Power Delay-Line Lengths

When the delay-line lengths need to be varied in real time, or interactively in a GUI, it is convenient to choose each delay-line length as an integer power of a distinct prime number [457]:

Suppose we are initially given desired delay-line lengths arranged in ascending order so that

*rounding*to the nearest integer (max 1). The prime-power delay-line length approximation is then of course

This prime-power length scheme is used to keep 16 delay lines both
variable and mutually prime in Faust's `reverb_designer.dsp`
programming example (via the function `prime_power_delays` in
`effect.lib`).

### Achieving Desired Reverberation Times

A lossless prototype reverberator, as in Fig.3.10 when , has all of its poles on the unit circle in the plane, and its reverberation time is infinity. To set the reverberation time to a desired value, we need to move the poles slightly inside the unit circle. Furthermore, due to air absorption (§2.3,§B.7.15), we want the high-frequency poles to be more damped than the low-frequency poles [314]. As discussed in §2.3, this type of transformation can be obtained using the substitution

where denotes the

*filtering per sample*in the propagation medium (a lowpass filter with gain not exceeding 1 at all frequencies).

^{4.14}Thus, to set the FDN reverberation time to at frequency , we want propagation through samples to result in attenuation by dB,

*i.e.*,

Solving for , the propagation attenuation per-sample, gives

The last form comes from ln, where denotes the time constant of decay (time to decay by ) [451],

*i.e.*,

Series expanding and assuming samples ( seconds) provides the practically useful approximation

#### Conformal Map Interpretation of Damping Substitution

The relation
[Eq.(3.7)] can
be written down directly from
[Eq.(3.5)] by interpreting Eq.(3.5) as an approximate
*conformal map* [326] which takes each pole
,
say, from the unit circle to the point
.
Thus, the new pole radius is approximately
,
where the approximation is valid when is approximately constant
between the new pole location and the unit circle. To see this,
consider the partial fraction expansion [449] of a proper
th-order lossless transfer function mapped to
:

Happily, while we may not know precisely where our poles have moved as a result of introducing the per-sample damping filter , the relation [Eq.(3.6)] remains exact at every frequency by construction, as it is based only on the physical interpretation of each unit delay as a propagation delay for a plane wave across one sampling interval , during which (zero-phase) filtering by is assumed (§2.3). More generally, we can design minimum-phase filters for which , and neglect the resulting phase dispersion.

In summary, we see that replacing by everywhere in the FDN lossless prototype (or any lossless LTI system for that matter) serves to move its poles away from the unit circle in the plane onto some contour inside the unit circle that provides the desired decay time at each frequency.

A general design guideline for artificial reverberation applications
[217] is that *all pole radii in the
reverberator should vary smoothly with frequency*. This translates
to having a *smooth frequency response*. To see why this
is desired, consider momentarily the frequency-independent case in
which we desire the same reverberation time at all frequencies
(Fig.3.10 with real , as drawn). In this case, it is
ideal for all of the poles to have this decay time. Otherwise, the
late decay of the impulse response will be dominated by the poles
having the largest magnitude, and it will be ``thinner'' than it was
at the beginning of the response when all poles were contributing to
the output. Only when all poles have the same magnitude will the late
response maintain the same modal density throughout the decay.

#### Damping Filters for Reverberation Delay Lines

In an FDN, such as the one shown in Fig.3.10, delays appear
in long delay-line chains . Therefore, the filter needed at
the output (or input) of the th delay line is
(replace
by
in Fig.3.10).^{4.15} However, because
is so close to in magnitude, and because it varies so weakly
across the frequency axis, we can design a much lower-order filter
that provides the desired attenuation
versus frequency to within psychoacoustic resolution. In fact,
perfectly nice reverberators can be designed in which is
merely *first order* for each
[314,217].

### Delay-Line Damping Filter Design

Let
denote the desired reverberation time at radian frequency
, and let denote the transfer function of the lowpass
filter to be placed in series with the th delay line which is
samples long. The problem we consider now is how to design these
filters to yield the desired reverberation time. We will specify an
*ideal* amplitude response for based on the desired
reverberation time at each frequency, and then use conventional
filter-design methods to obtain a low-order approximation to this
ideal specification.

In accordance with Eq.(3.6), the lowpass filter in series with a length delay line should approximate

This is the same formula derived by Jot [217] using a somewhat different approach.

Now that we have specified the ideal delay-line filter
in
terms of its amplitude response in dB, any number of filter-design
methods can be used to find a low-order which provides a good
approximation to satisfying Eq.(3.9). Examples include the functions
`invfreqz` and `stmcb` in Matlab. Since the variation
in reverberation time is typically very smooth with respect to
, the filters can be very low order.

#### First-Order Delay-Filter Design

The first-order case is very simple while enabling separate control of low-frequency and high-frequency reverberation times. For simplicity, let's specify and , denoting the desired decay-time at dc () and half the sampling rate ( ). Then we have determined the coefficients of a one-pole filter:

where denotes the th delay-line length in seconds. These two equations are readily solved to yield

The truncated series approximation

#### Orthogonalized First-Order Delay-Filter Design

In [217], first-order delay-line filters of the form

denotes the ratio of reverberation time at half the sampling rate divided by the reverberation time at dc.

^{4.16}

#### Multiband Delay-Filter Design

In §3.7.5, we derived first-order FDN delay-line filters which
can independently set the reverberation time at dc and at half the
sampling rate. However, perceptual studies indicate that
reverberation time should be independently adjustable in at least
three frequency bands [217]. To provide this degree
of control (and more), one can implement a multiband delay-line filter
using a general-purpose *filter bank*
[370,500]. The output, say, of each delay
line is split into bands, where is recommended, and then,
from Eq.(3.6), the gain in the th band for a length
delay-line can be set to

### Spectral Coloration Equalizer

In the previous section, a ``graphical equalizer'' was used to set the
reverberator decay time independently in each spectral band slice.
While this gives much control over decay time, there is no control
over the initial spectral gain in each band. Therefore, another good
place for a graphical equalizer is at the reverberator input or
output. Such an equalizer allows control of the *initial
spectral coloration* of the reverberator. In the example of
Fig.3.10, a spectral coloration equalizer is most efficiently
applied to the input signal , before entering the FDN (but after
splitting off the direct signal to be scaled by and added to the
output), or the output of in Fig.3.10.

### Tonal Correction Filter

Let denote the component of the impulse response arising from the th pole of the system. Then the energy associated with that pole is

*tonal correction filter*to be placed in series with the FDN, as shown in Fig.3.10.

In the case of the first-order delay-line filters discussed in §3.7.5, good tonal correction is given by the following one-zero filter:

### FDNs as Digital Waveguide Networks

As discussed in §C.15, the FDN using a Householder-reflection feedback matrix is equivalent to a network of digital waveguides intersecting at a single scattering junction [463,464,385]. The wave impedance in the th waveguide is simply , the th element of the axis-of-reflection vector . The choice corresponds to all of the waveguides having the same impedance (the ``isotrophic junction'' case).

### FDN Reverberators in Faust

The Faust example `reverb_designer.dsp` brings up a
FDN reverberator in which the signal out of each delay line is
split into five bands so that
can be controlled
independently in each band. The 16 delay-line lengths are distributed
exponentially between a minimum and maximum length set by two
min/max-length sliders, but rounded to the nearest integer-power of a
distinct prime, as introduced above in §3.7.3). The FDN
reverberator is implemented in Faust's `effect.lib`. The
band-splitting is carried out by the `filterbank` function in
Faust's `filter.lib`.

The Faust function `filterbank(order,freqs)` implements a
filter bank having the needed properties using Butterworth
lowpass/highpass band-splitting arranged in a dyadic tree (normally a
good choice for audio filter banks). That is, the whole spectrum is
split at the highest crossover frequency, the lowpass region is then
split into two bands at the next crossover frequency down, and so on,
splitting the lowpass band at each stage in the dyadic tree
[455,500]. The number of poles in each
Butterworth lowpass/highpass filter is specified by `order`,
and `freqs` contains a list of desired crossover frequencies
separating the bands. A certain amount of dispersion is also
introduced, since the filter bank is causal and delay-equalized (so
that the bands may be summed without phase cancellation artifacts at
the band edges). Also note that the lower bands are effectively
produced by higher order filters than the upper bands. When the
reverberation time is longer than the dispersion delay, the dispersion
should not be audible as such, although it can affect the ``sound'' of
the reverberation. In general, however, artificial reverberators
normally benefit from additional allpass dispersion.

Figure 3.12 shows the block diagram of a FDN
reverberator made from Faust's `reverb_designer.dsp` by
changing 16 to 4. Figure 3.13 shows the Faust block diagram of
the associated Hamard matrix multiplication. As it shows,
multiplication by a Hadamard matrix can be implemented (ignoring the
normalizing scale factor) as a series of block sums and differences
(often called *butterflies* or *shufflers*) in which the
block size decreases by a factor of 2 each stage. Figures for the
remaining components of the reverberator may be perused via the shell
command `faust2firefox reverb_designer.dsp` followed by
clicking on the blocks in the browser.

### Zita-Rev1

A FOSS^{4.17} reverberator
that combines elements of Schroeder (§3.5) and FDN
reverberators (§3.7) is
`zita-rev1`,^{4.18}written in C++ for Linux systems by Fons Adriaensen.
A Faust version of the `zita-rev1` stereo-mode functionality is
`zita_rev1` in Faust's `effect.lib`. A high-level
block diagram appears in Fig.3.14.

The main high-level addition relative to an 8th-order FDN reverberator
is the block labeled `allpass_combs` in Fig.3.14.
This block inserts a Schroeder allpass comb filter (Fig.2.30) in
series with each delay line. In `zita-rev1` (as of this
writing), the allpass-comb feedforward/feedback coefficients are all
set to . The delay-line lengths and other details are readily
found in the freely available source code (or by browsing the
Faust-generated block diagram).

#### Zita-Rev1 Delay-Line Filters

In `zita-rev1`, the damping filter for each delay line consists
of a low-shelf filter
[449],^{4.19}in series with a unique first-order lowpass filter that sets
the high-frequency to be half that of the middle-band at a
particular frequency (specified as ``HF Damping'' in the GUI).
Since the filter is constrained to be a lowpass,
for , *i.e.*, the decay time gets
shorter at higher frequencies.

Viewing the resulting damping filter
as a
three-band filter bank (§3.7.5), let and denote the
desired band gains at dc and ``middle frequencies'',
respectively.^{4.20} Then the low shelf may be set for a
desired dc-gain of , and its input (or output) signal
multiplied by to obtain the resulting filter

The lowpass filter is also first order, and to provide half the middle-band at the beginning of the ``high'' band, the lowpass should ``break'' to a gain of at the ``HF Damping'' frequency specified in the GUI. A unity-dc-gain one-pole lowpass has the form [449]

### Further Extensions

Schroeder's original structures for artificial reverberation were comb filters and allpass filters made from two comb filters. Since then, they have been upgraded to include specific early reflections and per-sample air-absorption filtering (Moorer, Schroeder), precisely specified frequency dependent reverberation time (Jot), and a nearly independent factorization of ``coloration'' and ``duration'' aspects (Jot). The evolution from comb filters to feedback delay networks (Gerzon, Stautner, Puckette, Jot) can be seen as a means for obtaining greater richness of feedback, so that the diffuseness of the impulse response is greater than what is possible with parallel and/or series comb filters. In fact, an FDN can be seen as a richly cross-coupled bank of feedback comb filters whenever the diagonal of the feedback matrix is nonzero. The question then becomes what aspects of artificial reverberation have not yet been fully addressed?

#### Spatialization of Reverberant Reflections

While we did not go into the subject here, the early reflections
should be *spatialized* by including a head-related transfer
function (HRTF) on each tap of the early-reflection delay line
[248].^{4.21}

Some kind of spatialization may be needed also for the late
reverberation. A true diffuse field (§3.2.1) consists of a
sum of plane waves traveling in *all* directions in 3D space.
Since we do not know how to achieve this effect using current systems
for reverberation, the typical goal is to simply extract
*uncorrelated* outputs from the reverberation network and feed
them to the various output channels, as discussed in §3.5.
However, this is not ideal, since the resulting sound field consists
of wavefronts arriving from each of the speakers, and it is possible
for the reverberation to sound like it is emanating from discrete
speaker locations. It may be that spatialization of some kind can
better fool the ear into believing the late reverberation is coming
from all directions.

#### Distribution of Mode Frequencies

Another way in which current reverberation systems are ``artificial''
is the unnaturally uniform distribution of resonant modes with respect
to frequency. Because Schroeder, FDN, and waveguide reverbs are
all essentially a collection of delay lines with feedback around
them, the modes tend to be distributed as the superposition of the
resonant modes of feedback comb filters. Since a feedback comb
filter has a nearly harmonic set of modes (see §2.6.2),
aggregates of comb filters tend to provide a *uniform* modal density in
the frequency domain. In real reverberant spaces, the mode density
increases as frequency squared, so it should be verified that the
uniform modes used in a reverberator are perceptually equivalent to
the increasingly dense modes in nature. Another aspect of perception
to consider is that frequency-domain perception of resonances actually
*decreases* with frequency. To summarize, in nature the modes get denser
with frequency, while in perception they are less resolved, and in
current reverberation systems they stay more or less uniform with
frequency; perhaps a uniform distribution is a good compromise between
nature and perception?

At low frequencies, however, resonant modes are accurately perceived
in reverberation as boosts, resonances, and cuts. They are analogous
to early reflections in the time domain, and we could call them the
``early resonances.'' It is interesting that no system for artificial
reverberation except waveguide mesh reverberation (of which the author
is aware) explicitly attempts precise shaping of the low-frequency
amplitude response of a desired reverberant space, at least not
directly. The low-frequency response is shaped indirectly by the
choice of early reflections, and the use of parallel comb-filter banks
in Schroeder reverberators serves also to shape the low-frequency
response significantly. However, it would be possible to add filters
for shaping more carefully the low-frequency response. Perhaps a
reason for this omission is that hall designers work very hard to
*eliminate* any explicit resonances or antiresonances in the
response of a room. If uneven resonance at low frequencies is always
considered a defect, then designing for a maximally uniform mode
distribution, as has been discussed for the high-frequency modes,
would be ideal also at low frequencies. Quite the opposite situation
exists when designing ``small-box reverberators'' to simulate musical
instrument resonators [428,203]; there,
the low-frequency modes impart a characteristic timbre on the low-frequency
resonance of the instrument (see Fig.3.2).

#### Digital Waveguide Reverberators

It was mentioned in §3.7.8 above that FDNs can be formulated as special cases of Digital Waveguide Networks (DWN) (see Appendix C for a fuller development of DWNs). Specifically, an FDN is obtained from a DWN consisting of a single scattering junction (§C.15). It follows that the DWN paradigm provides a more generalized framework in which to pursue further improvements of reverberation architecture. For example, when multiple FDNs are embedded within a single DWN, it becomes possible to richly cross-couple them in an energy-controlled manner in order to create richer recursive structures than either alone. General DWNs were proposed for artificial reverberation in [430,433].

#### The Digital Waveguide Mesh for Reverberation

A special case of digital waveguide networks known as the
*digital waveguide mesh*
has also been proposed for use in artificial reverberation systems
[396,518].

As discussed in §2.4, a digital waveguide (bidirectional delay
line) can be considered a computational acoustic model for traveling
waves in opposite directions. A *mesh* of such waveguides in 2D
or 3D can simulate waves traveling in *any* direction in the
space. As an analogy, consider a tennis racket in which a rectilinear
mesh of strings forms a pseudo-membrane.

A major advantage of the waveguide mesh for reverberation applications is that wavefronts are explicitly simulated in all directions, as in real reverberant spaces. Therefore, a true diffuse field can be developed in the late reverberation. Also, the echo density grows with time and the mode density grows with frequency in a natural manner for the 2D and 3D mesh. Finally, the low-frequency modes of the reverberant space can be simulated very precisely (for better or worse).

The computational cost of a waveguide mesh is made tractable relative
to more conventional finite-difference simulations by (1) the use of
multiply-free scattering junctions and (2) very coarse meshes. Use of
a coarse mesh means that the ``physical modeling'' aspects of the mesh
are only valid at low frequencies. As practical matter, this works
out well because the ear cannot hear mode tuning errors at high
frequencies. There is no error in the mode dampings in a lossless
reverberator prototype, because the waveguide mesh is lossless by
construction. Therefore, the only errors relative to an ideal
simulation of a lossless membrane or space are (1) mode tuning error,
and (2) finite band width (cut off at half the sampling rate). The
tuning error can be understood as due to *dispersion* of the
traveling waves in certain directions
[518,399]. Much progress has been
made on the problem of correcting this dispersion error in various
mesh geometries (rectilinear, triangular, tetrahedral, etc.)
[521,398,399].

See §C.14 for an introduction to the digital waveguide mesh and a few of its properties.

#### Time Varying Reverberators

In real rooms, thermal convention currents cause the propagation path
delays to vary over time [58]. Therefore, for greater
physical accuracy, the delay lines within a digital reverberator
should *vary* over time. From a more practical perspective, time
variation helps to break up and obscure unwanted repetition in the
late reverberation impulse response [430,104].

**Next Section:**

Delay-Line Interpolation

**Previous Section:**

Freeverb